Integrated RF front end with stacked transistor switch

ABSTRACT

A monolithic integrated circuit (IC), and method of manufacturing same, that includes all RF front end or transceiver elements for a portable communication device, including a power amplifier (PA), a matching, coupling and filtering network, and an antenna switch to couple the conditioned PA signal to an antenna. An output signal sensor senses at least a voltage amplitude of the signal switched by the antenna switch, and signals a PA control circuit to limit PA output power in response to excessive values of sensed output. Stacks of multiple FETs in series to operate as a switching device may be used for implementation of the RF front end, and the method and apparatus of such stacks are claimed as subcombinations. An iClass PA architecture is described that dissipatively terminates unwanted harmonics of the PA output signal. A preferred embodiment of the RF transceiver IC includes two distinct PA circuits, two distinct receive signal amplifier circuits, and a four-way antenna switch to selectably couple a single antenna connection to any one of the four circuits.

This application is a divisional of copending and commonly assigned U.S.patent application Ser. No. 11/501,125 filed Aug. 7, 2006 and entitled“Integrated RF Front End with Stacked Transistor Switch”; whichapplication is a continuation of commonly owned U.S. Pat. No. 7,088,971,issuing Aug. 8, 2006 (patent issuing from patent application Ser. No.11/158,597 and filed Jun. 22, 2005) and entitled “Integrated RF FrontEnd”; which patent is a continuation-in-part of commonly owned U.S. Pat.No. 7,248,120, issuing Jul. 24, 2007 (patent issuing from patentapplication Ser. No. 10/875,405 and filed Jun. 23, 2004) and entitled“Stacked Transistor Method and Apparatus”; and which application is alsorelated to the following commonly owned U.S. patent documents: U.S. Pat.No. 5,663,570, issued Sep. 2, 1997 and entitled “High-Frequency WirelessCommunication System on a Single Ultrathin Silicon On Sapphire Chip,”U.S. Pat. No. 6,804,502, issued Oct. 12, 2004 and entitled “SwitchCircuit and Method of Switching Radio Frequency Signals,” U.S. Pat. No.7,719,343, issued May 18, 2010 and entitled “Low Noise Charge PumpMethod and Apparatus;” and commonly-assigned, co-pending, and publishedU.S. patent application Ser. No. 12/799,583 filed Apr. 27, 2010 andentitled “Low Noise Charge Pump Method and Apparatus” (published Aug.26, 2010 as U.S. application publication number US-2010-0214010-A1); andthe entire contents of each of the above-cited U.S. patent documents arehereby incorporated herein in their entireties by reference.

BACKGROUND OF THE INVENTION

1. Field of the Invention

This invention relates broadly to integrated electronic circuits, andmore specifically to RF transceiver circuitry.

2. Related Art

Wireless communications devices, especially handheld devices, areundergoing sustained development. Perhaps more than any other factor,the extreme popularity of cellular mobile telephones has motivatedimprovements in efficiency, speed, size and cost-effectiveness for RFtransmission circuits in handheld devices. Enhancing the efficiency ofsuch circuits is highly desirable so that the size of the requiredbatteries may be reduced, while their life is extended.Cost-effectiveness is clearly always desirable for consumer products,particularly when such products require periodic replacement to stayabreast of changes in the technology. The steady advance offunctionality in cellular telephones, combined with consumer preferencesfor light and small devices, puts a premium on reducing the volumerequired for RF transmission circuits. Additionally, transmitters mustmeet stringent emission limits, which have been established in order tofacilitate high communication density at minimal power levels.

Most wireless communication units, such as cellular telephones, compriseat least one RF transceiver. A communication device, such as a cellulartelephone, may comprise a multiplicity of RF (radio frequency) front endcircuits, which are of primary interest herein. RF front end circuits(or subcircuits) typically include an RF transmit signal amplifier, aPower Amplifier (PA), a matching and filtering section, an antennaswitch, and may include a received signal amplifier. A completetransceiver generally also includes a low-noise amplifier for thereceived signal. Of these circuits, the PA subcircuit is typically themost power-consuming portion of such transmitters, and, also typically,is the source of the most significant unintended or “spurious”emissions. In order to extend battery life, to meet stringent spuriousemissions standards, and to minimize the cost of these high-volumeconsumer items, there is a need to improve the speed and efficiency,while reducing spurious emissions and manufacturing costs, for such PAsubcircuits. Due to their need to handle high power, the PA and antennaswitch subcircuits consume the most integrated circuit area.Manufacturing costs for integrated circuits are strongly dependent onthe amount of device area required for each circuit. Consequently,substantial reductions in the area required for the various RFtransceiver subsections will generally lead to commensurate reductionsin manufacturing costs for transceiver circuits.

A range of PA topologies have been developed, each having differentadvantages. For example, PAs of class A, B, C, D, E and F are well knownin the art. The primary amplifying devices in PAs of classes A-C aredesigned to operate in an “active” region of their operating range, thusintentionally conducting current while voltage is present across thedevice.

PAs of classes D, E and F attempt to reduce the power loss caused bysuch linear operation by employing amplifier devices as switches thatminimize operation in active regions, rather than as linear amplifiers.However, the pulse-type outputs from such amplifiers generally requireextensive filtering in order to establish a narrow-band sinusoidaloutput, as is typically required. While normal operation of PAs inclasses D-F does not intentionally cause drive element devices toconduct while voltage is present across the devices, even switcheddevices consume real power due to current flowing while voltage ispresent during finite switching periods. Moreover, compared to drivedevices in analog PAs operating at the same transmission centerfrequency, drive devices in class D-F switching circuits must oftenoperate at much higher frequencies. The higher frequency signals includesignificant energy at undesired frequencies, and such undesired signalenergies not only consume circuit power, but also require filtering tomeet emission limits.

Integration of devices is generally desirable in order to improvevarious features of the resulting product, such as operating frequencyand reliability, and may also reduce overall manufacturing costs, aswell as likely reducing the volume occupied by the circuits. FieldEffect Transistors (FETs) are extremely popular for both linearamplification and switching purposes in integrated circuits. However,integrated circuit (IC) FETs have a limited capability to withstandvoltage between any two nodes, including gate-source, gate-drain, anddrain-source node pairs. Such voltage withstand limitations mayparticularly impair the usefulness of IC FETs in high power switchingcircuits, in which inductive voltages may greatly exceed the supplyvoltage. As a particular example, the transmission output powercapability of an RF PA is highly dependent upon the amplitude of theoutput voltage. One of the difficulties with existing PA technologies isthat many otherwise desirably high-speed devices are fabricated usingprocesses that tend to yield FETs having relatively low breakdownvoltages. It is very desirable to solve this problem ant thereby providea wider voltage range while retaining other desirable integrated devicefeatures. Such a solution enables integration on monolithic integratedcircuits of power and control features that previously required separateprocessing, such as PA features and RF switch features. Integration ofinteracting circuits that were previously discrete will enhance yieldand predictability, due to the process matching that is inherent inmonolithic integration.

Methods and circuits are described herein that facilitate thefabrication of all of the transceiver RF circuits of a dual-bandtransceiver onto a single integrated circuit, thereby solving theproblems and gaining the benefits noted above. Many of the benefits areachieved by integrating even the front-end portions of transceivers thatdo not necessarily include dual-band operation. One or more alternativesare described for each of numerous subcircuits (or correspondingmethods), and a fully integrated RF front end, or an integrated RFtransceiver, may be fabricated by using any compatible one of suchalternatives for each section of the transceiver. Moreover, several ofthe subcircuits (or corresponding methods) that permit an integrated RFtransceiver to be realized are also useful in other contexts, oftenindependently of other RF transceiver subcircuits. Thus, varioussubcombinations of features described herein constitute usefulinventions in their own right. Combined, various aspects of thesesubcombinations together achieve an integrated dual-band RF transceiverhaving all of the benefits noted above. Particularly notable among theindependently useful subcircuits are stacked-FET RF switches andparticular PA circuit topologies. Finally, the integration of certain RFtransceiver subsections permits efficiencies in manufacturing withoutcompromising safety and reliability of the final product.

SUMMARY

A combination of methods and/or circuits is described that enables thefabrication of a self-protected monolithic integrated circuit includingall of the RF front-end sections of a communications transceiver. Suchself-protected RF front-end circuits particularly include thosesections, from a Power Amplifier (PA) through an antenna connection,that permit efficient internal protection from overload due to animproper, missing or damaged antenna.

Several subcombinations of the self-protected monolithic integrated RFfront-end circuits have independent importance. One such subcombinationis an integrated stacked-FET switch. One embodiment of thissubcombination is a circuit including a multiplicity of FETs in a stackhaving drain-source channels coupled in series to control conductivitybetween nodes in a circuit. A control signal is coupled to a first FETto cause changes in conductivity of the first FET, and conductivity ofthe remaining FETs is enslaved to the conductivity of the first FET. Avoltage withstand capability across the series combination of the FETstack may be substantially equal to a sum of drain-source voltagewithstand capabilities of the individual FETs of the stack. A gate ofeach FET other than the first FET may be capacitively coupled to acommon voltage.

Another subcombination is an RF Power Amplifier (PA) that may bereferred to as an integrated iClass PA. One embodiment of thissubcombination includes an input controlling an RF switch whose outputis coupled to a supply source via an RF choke, and which operates with acharacteristic drive output impedance at an operating frequency f₀. Thedrive output is coupled to an antenna connection having an expectedantenna impedance via a coupling circuit that matches the drive outputimpedance to the expected antenna impedance, and also includes a circuitthat dissipatively terminates signals at one or more harmonics of theoperating frequency f₀. The iClass RF PA may further include a shuntfilter configured to provide local minimum impedances between the driveoutput and the reference at a plurality of frequencies, including aneven harmonic of f₀ and a non-unity odd harmonic of f₀. These minimumimpedances may be approximately equal to the characteristic driveimpedance.

One embodiment of the self-protected front-end circuit is an integratedcircuit that includes a PA having an output amplitude regulator circuitthat is controlled by an output limiting controller. The embodimentfurther includes an antenna switch that selectably couples an antennaconnection to either the PA, via a matching and coupling circuit, or toa receive signal amplifier. The embodiment also includes an antennaconnection sensor configured to sense current through, and/or voltageat, the antenna connection, and circuitry coupling an output of theantenna connection sensor to the output limiting controller, which isconfigured to affect the output amplitude regulator circuit to preventthe existence of currents or voltages in excess of design limits at theantenna connection.

A related embodiment is a method of making a monolithically integratedPA with protection from excessive output values caused by high VoltageStanding Wave Ratios (VSWRs) that are due to improper antenna impedance.The embodiment includes fabricating an RF PA on an integrated circuitchip to receive a transmit signal, and providing an output powerlimiting circuit for the PA. It also includes fabricating coupling,matching and filtering circuits on the same integrated circuit tocondition a PA output signal having a PA output impedance to a differentimpedance desired for a connecting element that is connected to theintegrated circuit to couple the conditioned signal to an antenna. Theembodiment further includes disposing an antenna switch on theintegrated circuit between the PA and the connecting element whereby theconnecting element may be controllably coupled to either the conditionedsignal from the PA, or to a receive amplifying circuit disposed on theintegrated circuit. The embodiment includes providing a sensing circuitto sense a parameter of the signal delivered to the connecting element,and a PA control circuit to reduce power of the PA output signal inresponse to a value of the sensed parameter that is deemed excessive.

An embodiment of a further subcombination is a method of amplifying RFsignals, and includes providing a plural-FET stack to control conductionbetween an output drive node and a reference node to effect acharacteristic impedance at an operating frequency f₀. The embodimentfurther includes disposing, between the output drive node and thereference node, a shunt filter configured to dissipatively terminate aharmonic frequency of f₀. The shunt filter may include local minimumimpedances at an even harmonic of f₀ and at a non-unity odd harmonic off₀, and the local minimum impedances may be approximately equal to thecharacteristic impedance of the FET stack.

BRIEF DESCRIPTION OF THE DRAWINGS

Embodiments of the present invention will be more readily understood byreference to the following figures, in which like reference numbers anddesignations indicate like elements.

FIG. 1 is a block schematic diagram representative of some types of RFpower amplifier (PA) circuitry.

FIG. 2 is a generalized schematic diagram of a stacked-FET controlcircuit useable with a PA such as shown in FIG. 1.

FIG. 3 is a simplified schematic diagram of an impedance matching andcoupling bandpass filter useable in a PA such as shown in FIG. 1.

FIG. 4 is a simplified schematic diagram of a shunt filter useable in aPA such as shown in FIG. 1.

FIG. 5 is a simplified schematic diagram of a shunt power controlcircuit useable in a PA such as shown in FIG. 1.

FIG. 6 is a simplified schematic diagram of an output filter for Class Foperation of a PA such as shown in FIG. 1.

FIG. 7 is a schematic diagram illustrating alternative biasing featuresfor a stacked-FET amplifying circuit.

FIG. 8 is a simplified schematic diagram illustrating an alternativemethod of biasing FETs of a FET stack.

FIG. 9 is a simplified schematic diagram of a Class D PA employingcomplementary stacked-FET drive elements.

FIG. 10 is a schematic diagram of exemplary output filtering for a ClassD PA such as represented in FIG. 9.

FIG. 11 is a simplified schematic diagram of a harmonic terminationshunt filter useable in a PA such as shown in FIG. 1.

FIG. 12 is a simplified schematic diagram of an integrated dual-band RFtransceiver.

FIG. 13 is a schematic diagram of an exemplary PA output powersupervisor for directing output power levels in the transceiver of FIG.12.

FIG. 14 is a schematic diagram of an exemplary series output regulatorfor the transceiver of FIG. 12.

FIG. 15 is a schematic diagram of an exemplary pulse width controllerfor the transceiver of FIG. 12.

FIG. 16 is a schematic diagram of an exemplary output voltage detectorfor the transceiver of FIG. 12.

FIG. 17 is a schematic diagram of an exemplary dual-band antenna switchfor the transceiver of FIG. 12.

DETAILED DESCRIPTION

I. Power Amplifier Overview

FIG. 1 is a block diagram of an RF power amplifier (PA). The illustratedRF PA is quite general, in that varying the biasing of devices in adriver elements block 200, and/or varying details of the other blocks,will permit the RF PA illustrated in FIG. 1 to operate in any ofamplifier classes A, B, C, E, F, or, as described further herein, as aniClass amplifier. FIGS. 2-5 each show an exemplary circuit to implementone of the blocks shown in FIG. 1.

An input 102 is provided to the PA with respect to a circuit reference,or common, 104. The input 102 generally comprises a properly biasedsignal at a center drive frequency, f₀. In response to the input 102,the driver elements block 200 controls conduction between a drive outputnode 106 and the circuit common 104. The driver elements block 200, inconjunction with current from V_(DD) via an RF choke (RFC) L_(S) 108,provides a signal having a particular impedance Zdrive. Zdrive may varywith frequency, but will refer to the drive impedance at the centeroperating frequency f₀, unless otherwise indicated. A shunt filter 400may be coupled between the drive output node 106 and the circuit common104. Numerous different filtering arrangements may be used, someexamples of which are described subsequently herein.

An antenna 110 has a characteristic impedance Z_(OUT), generally 50Ω (atthe center frequency f₀ unless otherwise indicated). A block 300 istypically required to provide matching and coupling between the drivenode 106 (at Zdrive) and the output at Z_(OUT). Following the matchingand coupling, an output filter section such as the combination of L_(O)116 and C_(O) 118 may typically be disposed in the signal path before anRF switch, S_(RF) 120, which appropriately couples the output to theantenna 110. Because the PA circuit is integrated on a semiconductordevice, and the antenna 110 is typically external to the IC comprisingthe PA, the antenna 110 often operates with a different referencevoltage, for example a chassis ground 112, which has a non-zeroimpedance to the circuit common 104. Accordingly, the matching-couplingblock 300, as well as the filter section 116-118, has an output that isreferenced to chassis ground 112.

Power control may optionally be provided. One example employs a shuntpower control block 500, which may provide a voltage offset betweenchassis ground 112 and circuit common 104 to reduce the amplitude ofsignals received by the antenna 110. A series regulator circuit, such asitems 1400-1401 in FIG. 12, is probably used more commonly.

The monolithically integrated RF PAs, RF front ends, and RF transceiversdescribed herein may be fabricated to operate at relatively highfrequencies of at least 900 MHz and/or 2.4 GHz, and at moderate powerlevels. These designs are useful for transceivers having transmit powermaximums of at least 0.5 W, 1 W, or 1.5 W RMS of RF output powerdelivered to the antenna connection when it is properly coupled to amatched antenna.

II. Stacked-FET Drivers

FIG. 2 is a simplified schematic diagram of a stacked-FET circuit thatmay be used for the driver elements block 200 in the RF PA of FIG. 1,for controlling conduction between the drive output node 106 and thecircuit common 104. The stack includes two or more FETs of the samepolarity, i.e., all FETs in a stack are N-channel FETs, or all areP-channel FETs, or at least all FETs in a stack operate substantiallysimilarly as each other.

The FET stack 200 of FIG. 2 is configured to control conduction betweentwo nodes of an integrated circuit. A terminal Vdrive_(REF) 202 isconnected to one of the two nodes (e.g., circuit common 104 in FIG. 1),while a terminal Vdrive 224 is connected to the other node (e.g., Vdrive106 in FIG. 1). For N-channel FETs (N-FETs) as illustrated in FIG. 2,Vdrive_(REF) 202 will be connected to the more negative of the twonodes, for example to circuit common 104 in FIG. 1. The terminalVdrive_(REF) 202 is coupled to the source of a first FET of the stack200, M₁ 204.

The FET stack 200 is controlled by means of an input signal, relative toterminal Vdrive_(REF) 202, that is coupled to the gate of thesignal-input FET M₁ 204 via an input terminal 206. The drain of M₁ 204is coupled to the source of a second FET M₂ 208. The gate of M₂ 208 isprovided with a bias voltage VB₂ 210 via a bias resistor RB₂ 212, and isdecoupled to Vdrive_(REF) 202 via CG₂ 214. In some embodiments, thesetwo FETs are sufficient, when properly configured to divide appliedvoltages so as to avoid exceeding breakdown limits of either device, toserve as a conduction controlling circuit to handle increased voltagesin a circuit such as a PA or a quad mixer.

In other embodiments, however, one or more additional FETs of the samepolarity are connected in series with M₁ 204 and M₂ 208. Such additionalFETs are represented in FIG. 2 by an Nth FET, M_(N) 216. As for eachadditional FET of the stack, the source of M_(N) 216 is coupled to thedrain of the preceding FET of the stack, i.e., to the drain of FETM_(N-1) (not shown, though if N=3 then M_(N-1) is M₂ 208). The drain ofthe last FET of the stack, M_(N) 216, is coupled to the output terminalVdrive 224. Associated with each additional FET is a biasing voltageVB_(N) 218, which is coupled to the gate of the FET via a bias impedancesuch as RB_(N) 220, and a capacitor CG_(N) 222 for coupling the gate toa voltage such that the FET is enslaved to conduction by thesignal-input FET (here, M₁ 204). As shown, enslaving may be effected bycoupling the gate of each additional FET to Vdrive_(REF) 202.

FET stacks with at least nine FETs in series have been fabricated orsimulated, and stacks of even more series FETs are certainly possible.Note that physical circuit couplings generally include finitecapacitance, inductance, and resistance. For many purposes it ispreferred that the FETs of the FET stack 200 be coupled with minimalimpedance in series, drain to source. However, impedance may beintentionally added to such couplings. For example, it may be desirableto more closely control a drive impedance, and to dissipate heat inspecific resistive series coupling elements rather than within the FETsthemselves. It may also be desirable to add impedance between the FETsof the FET stack 200 so as to tune the conductance of the drive circuit.

II.A. FET Stack Biasing

In some embodiments, the FETs of a FET stack may all have substantiallysimilar voltage withstand capabilities, such as breakdown voltagesV_(GS(br)), V_(DS(br)), and V_(DG(br)). For some integrated circuitfabrication processes, these values will be similar from FET to FET.Moreover, for some integrated circuit fabrication processes, thebreakdown voltages V_(GS(br)), V_(DS(br)), and V_(DG(br)) may beapproximately equal to each other. Proper biasing will usefully ensurethat none of these breakdown voltages is exceeded during normaloperation of the circuit. In some embodiments, with proper biasing,voltage excursions between Vdrive_(REF) 202 and Vdrive 224 may bepermitted to approach a sum of V_(DS) breakdown voltages for eachconstituent FET of the stack.

Biasing and coupling the FETs of a FET stack as described below mayprevent voltages from exceeding any maximum allowable node to nodevoltage for any FET of the stack, even when the total voltage impressedfrom Vdrive 224 to Vdrive_(REF) 202 is nearly equal to the sum of themaximum allowable V_(DS) for the individual FETs of the stack. Unlessotherwise noted, the maximum allowable voltage between any two nodes ofthe FETs (i.e., V_(GS), V_(DS), and V_(DG)) are generally assumed to besubstantially equal, both for the various nodes of each FET, and fromFET to FET, which accords with an exemplary semiconductor fabricationprocesses. However, the skilled person may readily extend the principlesset forth below to encompass situations in which these maximum allowablevoltages are not equal. Also, the calculations set forth below forN-channel FET stacks may be applied to P-channel FET stacks withappropriate inversions of polarities and references.

The impedance of the gate drive of M₁ may be selected according toordinary transistor driving principles. In this exemplary embodiment,V_(DS)(max) is the same for all FETs of the stack. V_(DS) for M₁ willtherefore approximate (Vdrive-Vdrive_(REF))/N. For each FET M_(“X”), forX values from 2 to N, the effective value of each biasing resistorRB_(X) is selected to control a time constant, τ_(GX), of the gatecoupling. τ_(GX) is, approximately, the sum of effective capacitances ofthe gate coupling capacitor CG_(X) plus the parasitic gate capacitancesC_(GPX), multiplied by the series impedance to a biasing voltage. Suchseries impedance is typically resistive, and will be designatedRB_(X(equiv)). It may be desirable for τ_(GX) to be much longer than theperiod 1/f₀ of the center drive frequency, preferably 5-20 times aslong. Thus, a good design center goal is:RB _(X(equiv))(C _(GX) +C _(GPX))=10/f ₀  (Eqn. 1).

With respect to Vdrive_(REF), and for Vpeak that is the maximum expectedvalue of Vdrive, one proper bias voltage value is simply a proportionalportion of ½ of Vpeak:VB _(X) =X(Vpeak)/2N  (Eqn. 2)Thus, an example in which N=4 yields: VB₂=Vpeak/4, VB₃=3(Vpeak)/8, andVB₄=Vpeak/2.II.B. FET Stack Gate Signal Coupling

In FIGS. 2 and 7, each gate node (V_(GX)) is coupled via the capacitorC_(GX) to the reference voltage Vdrive_(REF). Each gate node is alsocoupled to a DC bias voltage via a bias resistor. In this configuration,the effective drive voltage V_(GSX) for each FET M_(X) of the FET stackdepends upon the voltage excursion of its source, V_(SX), in conjunctionwith the impedance from the source to the gate node, and from the gatenode to AC ground. These impedances are dominated by the gate-sourcecapacitance and the coupling capacitor CG_(X). Appropriate values forCG_(X) may be determined as follows.

In the exemplary embodiment, the maximum voltage between each node pairof each FET is the same. The voltage excursions of the source of FET M₂must therefore not exceed the maximum V_(DS) for M₁. As such, the valueof CG₂ is unlimited, and desirably large, for effecting AC grounding ofthe gate of M₂ and thereby providing the largest common-gate drivesignal to M₂. V_(GS) (max) will not be exceeded for M2 if the (DC)voltage on the gate is maintained within the range of the source voltageexcursions. However, if (contrary to the assumptions above) the maximumV_(DS1) exceeds the maximum V_(GS2), then CG₂ values may need to belimited in a manner analogous to that described below for CG_(X) for Xfrom 2 to N.

The voltage excursion of the source of each FET M_(X) with respect toVdrive_(REF), ΔV_(SX), will be equal to the drain voltage excursion forM_((X-1)), ΔV_(D(X-1)). This voltage, presuming equal division betweenthe various FETs, is X(Vpeak−Vmin)/N. For Vmin=0, this is simplyX(Vpeak)/N, and ΔV_(SX)=(X−1)(Vpeak)/N.

The parasitic gate-source capacitance C_(GS) of a FET increases, whenV_(GS)=V_(GS)(on), to C_(OX), the oxide capacitance. C_(OX) for aparticular FET M_(X) is designated C_(OXX). Because CG_(X) is coupled tothe reference voltage Vdrive_(REF), the net V_(GSX) will be capacitivelydivided between CG_(X) and C_(OXX). Thus, the gate-source excursionΔV_(GSX)=(ΔV_(SX))/(1+C_(OXX)/C_(GSX)). Presuming equal maximums forV_(GS) and V_(DS), it is desired to limit ΔV_(GSX)≦Vpeak/N. Thus,substituting for ΔV_(GSX) and ΔV_(SX),Vpeak/N≧[(X−1)(Vpeak)/N]/[1+C_(OXX)/C_(GSX)]. Appropriate consolidationyields:C _(GX) ≦C _(OXX)/(X−2)  (Eqn. 3)For X=2, C_(GX)≦infinity, as expected. Also as expected, excessivevalues for C_(GX) will tend to cause excessive gate-source voltageexcursions (ΔV_(GSX)). The inequality of Eqn. 3 may prevent excessivevoltages between nodes of the devices. However, C_(GX) may desirably beas large as is allowable so as to provide the largest allowable drivelevels without exceeding breakdown voltages. Accordingly, the inequalityof equation 3 may be treated as an approximate equality.

The result set forth above may not apply when it is desired to dividevoltage differently between different FETs of the stack, or when maximumgate-source voltages differ from maximum drain-source voltages. However,the skilled person will have no difficulty determining desirable valuesfor C_(GX) for such various circumstances by calculations correspondingto those set forth above, with appropriately modified assumptions.Because the capacitors C_(GX) must sustain voltages exceeding the biasvoltage of the corresponding FET M_(X), a metal-insulator-metal (MIM)capacitor is a good choice. Moreover, the capacitance of both(parasitic) oxide capacitors and MIM capacitors is a direct function ofgeometry. Certain fabrication variables, such as lithographic variables,therefore tend to have similar effects on both types of capacitances,leaving the ratio of such capacitances relatively immune to suchvariables.

FIG. 3 shows an exemplary matching, coupling and filtering block 300,which, as shown in FIG. 1, may be disposed between the drive output node106 and the antenna 110. The matching function transforms the typically50Ω characteristic impedance of the antenna, Z_(OUT), to thecharacteristic impedance of the drive output node 106 (both at theoperating frequency f₀) in a manner well known to those of skill in theart. The coupling capacitor C_(C) 302 blocks DC from the drive outputnode 106, and may be selected to have an impedance at f₀ that is less,and desirably much less, than the impedance to chassis ground 112, or tocircuit common 104, from either side of the capacitor 302. The matchingcircuit 300 includes an “A” filter element comprising an inductor L_(A)304 and a capacitor C_(A) 306, which may be fabricated as part of theintegrated circuit. The matching circuit 300 also includes a “B” filterelement comprising an inductor L_(B) 308 and a capacitor C_(B) 310 tochassis ground 112 (to which the antenna 110 of FIG. 1 is referenced).The coupling capacitor C_(C) 302, as well as the inductor L_(B) 308 andthe capacitor C_(B) 310 may be fabricated on an integrated circuit withthe PA, but some of these devices are typically external to theintegrated circuit.

III. Shunt Filtering

FIG. 4 illustrates a shunt filter 400 that may be employed in an iClassPA such as illustrated in FIG. 1. A node 402 of shunt filter 400 may beconnected to the drive output node 106 of FIG. 1, and an opposite node404 may be connected to circuit common 104 of FIG. 1. The shunt filter400 may provide local minimum impedances at each of one or moreparticular frequencies. The minimum impedances may be matched to thedrive circuit impedance (as established, for example, by the driveelements 200 and the RF choke L_(S) 108). The shunt filter elements maybe fabricated as part of the integrated circuit that includes the driveelements 200, thus reducing loop areas of currents passing through theelements of the shunt filter 400.

A shunt filter 400 for FIG. 1 may be a “transmission line filter”fabricated of reactive elements that are substantially distributed overan appropriate length, for example ¼ wavelength at f₀. Such atransmission line may be coupled to circuit common via a resonantcircuit having a maximum impedance at f₀, such as a filter 600 asillustrated in FIG. 6 and described in more detail hereinbelow. Such aconfiguration for the shunt filter 400 provides local impedance minimums(approximately zero) at each even harmonic of f₀, and local maximumimpedances at each odd harmonic of f₀. Stated more concisely, such aconfiguration may typically be said to reflect all odd harmonics, and toshort all even harmonics, thus permitting operation as a Class F PA.

However, the shunt filter 400 illustrated in FIG. 4 for use in an iClassPA generally differs from such a transmission line filter. First, theshunt filter 400 may employ lumped, rather than distributed, elements.Consequently, local minimum impedances may occur at selected frequenciesrather than at all odd, or at all even, harmonics of a resonantfrequency (e.g., f₀). Second, the filter may employ series resistiveelements to intentionally establish a non-zero value of local minimumimpedance. It may be useful, for example, to control the local impedanceminimum values so as to match an impedance of the drive circuit at thecorresponding frequency (or, alternatively, at f₀). As a result of suchdifferences between the circuit illustrated in FIG. 4 and a conventionaltransmission line filter, the magnitudes of currents in current loopsmay be reduced. Moreover, drive element power dissipation may be reducedat the frequencies corresponding to the selected minimum impedances.

In FIG. 4, a first shunt filter element includes L_(SF1) 406, R_(SF1)408, and C_(SF1) 410. These components establish a local minimumimpedance at a particular frequency, with the impedance increasing forboth higher and lower frequencies. A switch S1 412, in conjunction withan additional capacitor C_(SF3) 414, represents an optional circuit foradjusting the frequency of the minimum impedance of the first filterelement. As shown, the effective value of the series capacitor of thefirst filter element is increased when S1 is closed and C_(SF3) isdisposed in parallel with C_(SF1) 410.

Of course, such frequency adjustability may be effected in numerousdifferent manners. For example, S1 412 may be a FET for electronicallyswitching the frequency. Additionally or alternatively, C_(SF1) 410, aswell as optional C_(SF3) 414, may be varactors (with the correspondingaddition of an appropriate control circuit for the DC voltages on suchvaractors). Moreover, the capacitor C_(SF3) 414 may be disposed inseries connection, rather than parallel connection, with C_(SF1) 410, inwhich event the switch S1 may be configured to bypass the capacitorC_(SF3) 414. Yet further, analogous techniques may be employed to varyinductance, rather than capacitance. For example, the switch S1 412 mayselectably bypass a second inductive element, so as to vary theeffective inductance of the first shunt filter element.

The second shunt filter element comprises an inductor L_(SF2) 416, aresistive element R_(SF2) 418, and a capacitor C_(SF2) 420. The resonantfrequency of the second filter element (or, indeed, of any furtherfilter element) of the shunt filter 400 may be varied by similartechniques as outlined above with respect to the first filter element.It may be useful to have a minimum impedance that is substantiallyresistive, and/or that is non-zero. In one embodiment, the first andsecond filter elements are designed to provide local minimum impedances,at a second harmonic and a third harmonic of the operating frequency f₀respectively, which are approximately equal to the drive circuitimpedance. Though only two filter elements are illustrated, additionalharmonics may desirably be treated with additional filter elements (notshown).

FIG. 11 is a schematic diagram of a circuit alternative for shunt filter400 together with matching and coupling circuit 300. Zdrive and Z_(OUT)of FIG. 11 are connected as shown in FIG. 1. C_(C) 302 is substantiallythe same as in FIG. 3, providing DC isolation of the PA from the antennaoutput. The shunt filter includes a parallel resonant circuit primarilyconsisting of C_(SF) 980 and L_(SF) 982, which together function as atank circuit that is resonant at f₀. All integer harmonic frequencies off₀ are coupled through Rdrive 984, which is preferably selected to beapproximately equal to the characteristic drive impedance of the PAswitching circuit. Thereby, all harmonic frequencies of f₀ areterminated at the drive impedance. In some embodiments, alternativefiltering, such as two parallel tank circuits each resonant near f₀, maybe disposed in series above Rdrive 984. Dual tank circuits may beconfigured either to resonate at substantially identical frequencies,thereby increasing the impedance at f₀ and reducing power loss at f₀, orto resonate at slightly different frequencies, thereby broadening therange of frequencies at which the circuit has high impedance to easemanufacturing tolerances. Following the shunt filter, a matching andfiltering network 990 may be as illustrated in FIG. 3, except for twodifferences: first, coupling capacitor C_(C) 302 is omitted, and second,C_(A) 306, being on the antenna side of coupling capacitor C_(C) 302, iscoupled to earth ground 112 rather than circuit common 104. The twocommon references may be made effectively identical in some integratedcircuit layouts.

IV. PA Output Power Control

FIG. 5 illustrates elements of one possible shunt power control circuit500 for the PA of FIG. 1. A power control input 502 may establish a biaswith respect to chassis ground 112. An inductive impedance Lg1 504(which may reflect, for example, the inductance of a bond wire) isillustrated between chassis ground and the source of a power control FETM_(PC) 506. A connection 508 may be coupled to circuit common 104 inFIG. 1. An inductance Lg2 510 typically exists in series with a bypasscapacitor for the power control circuit, C_(PC) 512. Assuming that theDC voltage V_(DD) is with respect to chassis ground 112, thesubstantially DC voltage established across C_(PC) 512 will reduce theeffective supply voltage with respect to circuit common 104.

Other techniques may also be used to control the output power for acircuit as shown in FIG. 1, particularly when operated in a Class Fconfiguration (rectangular wave input control). Such other techniquesmay be used either in addition to, or instead of, shunt power control asdescribed above with respect to FIG. 5.

As a first example, referring also to FIG. 2, the bias voltages on theFETs M₂ 208 . . . M_(N) 216 may be adjusted. Efficiency will decrease,but power output will decrease more rapidly. As noted above, bias maygenerally be set such that VB_(X)=X(Vpeak)/2N. However, if VB2 isdecreased well below the calculated value, the output voltage Vdrive (iniClass operation with harmonic termination) will also decline. Thus, forexample, a circuit may be configured as shown in FIG. 7, except that theeffective average voltage at the gate of M₂ 208 may be controllablyreduced. This may be accomplished by making the value of RB₁ 708variable (e.g., by means of a parallel FET). Alternatively, the value ofRB₁ 708 may be reduced, and RB₁ 708 may be coupled to a variable voltagesource rather than to reference 202. Reducing the bias voltage willcause corresponding reductions in drive output voltages. As describedbelow in more detail, the self adjusting bias supply circuit of FIG. 7will permit the bias on RB_(N) to gradually follow the reductions inVdrive(peak) that are caused by varying the bias voltage on the gate ofM₂.

PA output power may also be controlled by varying the amplitude of thedrive signal. The conduction impedance of the drive elements will behigher when driven with a lower amplitude rectangular wave, resulting ina smaller effective drive voltage. The efficiency of this technique istypically comparable to the efficiency of varying bias voltages.

As discussed below with respect to FIGS. 12 and 14, a series regulatorcircuit may be used to control PA output power either alone, or inconjunction with one or more other power control techniques.

V. Alternative PA Embodiments

FIG. 6 is a simplified schematic diagram of a filter circuit 600 thatmay be employed in a manner similar to the shunt filter 400 in FIG. 1 toform a versatile PA architecture. The drive output node 106 of FIG. 1may be coupled to a Zdrive node as an input to 600. The Zdrive node maybe coupled via a ¼ wavelength transmission line 602 and a couplingcapacitor C_(C1) 612 to an output filter section. The output filtersection may comprise a parallel combination of L_(OF1) 610 and C_(OF1)608, resonant at the operating frequency f₀. Unlike some embodiments ofshunt filters 400, the output filter section of FIG. 6 is typically notpart of the PA integrated circuit, and thus is referenced to chassisground 112 rather than to circuit common. The impedance of this bandpassfilter to ground 112 falls rapidly as the frequency deviates from f₀,and, therefore, the harmonics of the operating frequency are effectivelyshorted to ground at the output filter end of the transmission line 602.The standing waves of the properly tuned ¼ wavelength transmission linetherefore provide a high impedance at each odd harmonic, and a lowimpedance at each even harmonic, as seen at the Zdrive node. A Z_(OUT)node 604 may be coupled to a further output filter section 116-118, anRF switch 120, and antenna 110 as shown in FIG. 1. A matching network(not shown) may also be required, similar to that illustrated in FIG. 3with the coupling capacitor C_(C) 302 omitted. Such further filteringand matching circuits, or a transmission line coupled thereto, willideally appear to the filter circuit 600 as a resistive impedanceR_(OUTequiv) 606 at the operating frequency f₀.

Modified as described above, the circuit of FIG. 1 may be operated as anRF PA of Class A, Class B, Class C, Class E or Class F. For Class Aoperation, the input signal 102 is sinusoidal and does not cause thecurrent through M₁ 202 of FIG. 2 to go to zero. For Class B operation,the input signal 102 is sinusoidal but M₁ 202 conducts only 50% of thetime (conduction angle 180 degrees). Operation may be Class C, with aconduction angle less than 180 degrees, which yields some efficiencyimprovement as compared to the Class B operation. In each case the FETsM₂ to M_(N) are enslaved to M₁, and the FET stack of FIG. 2 functionssubstantially as a single device. The circuit of FIG. 1 may also beoperated as an iClass PA in a configuration related to Class F buthaving dissipative termination for harmonics of the operating frequency.

The circuit of FIG. 1, configured as described immediately above, mayalso be operated as a Class F RF PA. For Class F operation the inputsignal is preferably a square wave having a duty cycle that causes thecircuit 200 to conduct at precisely a 50% duty cycle. The outputvoltages resulting from Class F operation generally increasesubstantially when the conduction duty cycle deviates from 50%.Unfortunately, ordinary manufacturing component variations tend to causethe duty cycle to deviate from 50%, and consequently the circuit may notreadily be able to safely and reliably utilize the full voltagewithstand capability of the drive element(s).

VI. Alternative Bias and Slaving

Embodiments of a FET stack, as described herein, may include asignal-input FET that receives a drive signal coupled to its gate withrespect to a reference voltage that is coupled to its source. Theremaining FETs of the stack may be enslaved to the signal-input FET,such that they conduct under the control of conduction in thesignal-input FET. The method by which the other FETs of a FET stack areenslaved to the signal-input FET must cooperate with the method employedto properly bias the FETs. Accordingly, enslavement and biasing areaddressed together.

In RF PAs generally according to FIG. 1, the peak voltage of driveoutput node 106 (with respect to circuit common 104) will often exceedtwice the available supply voltage V_(DD) 114. As such, bias voltages asrequired for the driver elements of FIG. 2 may not be readily available.This lack may be remedied by recourse, for example, to a charge pump. Acharge pump that is preferred from the standpoint of minimal noisegeneration is described in commonly owned and copending U.S. patentapplication Ser. No. 10/658,154, “Low-Noise Charge Pump Method andApparatus,” which is hereby incorporated in its entirety by reference.As described therein in detail, a low-current voltage source may bereadily established at any desired voltage. Such voltage source may beprovided, as needed, to any of the bias voltage inputs VB₂ 210 to VB_(N)218 in FIG. 2.

FIG. 7 illustrates a self-adjusting bias supply that may be employed tobias the FETs of a FET stack. As in FIG. 2, a signal input 206 iscoupled to the gate of a signal-input FET M₁ 204. The source of the FETM₁ 204 is coupled to Vd_(REF) 202, while its drain is coupled in serieswith each subsequent FET of the stack, including M₂ 208 . . . M_(N) 216.The drain of the last FET of the stack, M_(N) 216, is coupled to Vdrive224. To provide a bias voltage that reflects Vdrive 224, a diode (orequivalent) D_(B) 702 charges a bias supply capacitor C_(B) 704 to Vbias706. Vbias will charge to approximately Vpeak, the peak value of Vdrive224 with respect to Vd_(REF) 202. If a time constant associated withC_(B) 704 is sufficiently long, then Vbias will remain substantially atthis value. The time constant is the product of the capacitance of C_(B)704 multiplied by the resistance, to Vd_(REF) 202, of the resistivevoltage divider having N resistors including RB₁ 708, RB₂ 710, . . . ,and RB_(N) 712. The total resistance of this voltage divider may bedesignated R_(B)sum.

With respect to Equations 1, 2 and 3 that are set forth above, “X”represents the position of the particular FET in a stack, and Nrepresents the total number of FETs in such stack. Assuming that allFETs are approximately identical, it may be seen that:RB₁=RB₂= . . . =RB_((N−1))  (Eqn. 4),and, accordingly,RB _(N)=(N−1)RB ₁  (Eqn. 5).

In view of equations 1-5, it may be seen that, for the last FET of thestack (X=N),(C _(GX) +C _(OXX))=C _(OX)(N−1)/(N−2)  (Eqn. 6),RB _(X(equiv)) =RB ₁(N−1)/2  (Eqn. 7),andRB ₁≧20(N−2)/[C _(OX)(N−1)² f ₀]  (Eqn. 8).Thus, for N=3, RB₁≧5/C_(OX)/f₀, and RB₁ declines monotonically as Nincreases (for given values of C_(OX) & f₀).

The total resistance R_(B)sum of the resistive divider described above,in which the lower (N−1) resistors are RB₁ and the top (or Nth) resistoris the sum of the lower resistors, is simply 2(N−1)RB₁. The ripple onVbias 706 may be acceptably low if the time constantC_(B)(R_(B)sum)≧10/f₀. Coupling that criteria with Eqn. 8 yieldsC _(B) ≧C _(OX)(N−1)/(N−2)/4  (Eqn. 9).Thus, for N=3, C_(B)≧C_(OX)/2. As N increases, smaller values of C_(B)(with respect to C_(OX)) will be required to achieve the same ripplevoltage.

A significant ripple voltage is not necessarily a problem, and C_(B) maydesirably assume even smaller values if rapid self-adjustment responseis required. Indeed, in view of the filtering effected by each gatebypass capacitor CG_(X) in conjunction with RB_(X(equiv)), an averagevalue is the main consideration for Vbias. However, if the average valueof Vbias is permitted to decline significantly below Vpeak for anyreason, including the presence of substantial ripple on C_(B), theskilled person will understand that the resistive divider values shouldbe adjusted accordingly.

FIG. 8 illustrates an alternative for providing both bias and gatecoupling for FETs M₃ and above (X≧3). A reference 802 is coupled to thesource of a signal-input FET M₁ 804, the gate of which is coupled to aninput signal 806. The drain of M₁ 804 is coupled to the source of asecond FET M₂ 808. A bias voltage is applied to a bias input 810, whichis coupled via a bias resistance RB 812 to the gate of M₂ 808, and to arelatively large capacitance CG₂ 814. The drain of M₂ 808 is coupled tothe source of a third FET of the stack, M₃ 816. The drain of M₃ 816 maybe coupled to a further FET stage, if present. However, the drain of theFET of the last stage, M₃ 816 as shown in FIG. 8, is coupled to anoutput node Vdrive 818.

The gate 820 of FET M₃ 816 may be coupled to the base of the precedingstage FET M₂ 808 via a zener diode DZ 822. DZ 822 may have a conductionthreshold knee at approximately the maximum desired value for V_(DS) ofM₃ 816. (A circuit operating similarly to a zener diode may be usedinstead of DZ 822.) Additional FET stages designated by subscripts “Y”may be added. For such additional stages, corresponding additional zenerdiodes may be employed in like manner as DZ 822, i.e., anode to the gateof additional FET M_(Y), and cathode to the gate of M_((Y-1)).

VI.A. Alternative Stacked FET Switch Configurations and Extensions

The FET stacks described above with respect to FIGS. 1-8 employN-channel FETs (N-FETs). P-channel FET (P-FET) stacks may be fabricatedanalogously, by reversing the polarity of each voltage and polarizedcomponent associated with the stack. The P-FET stack reference voltagewill generally be coupled to the source of a first, signal-input FETM_(P1). Such inverted circuits will operate according to substantiallythe same principles as the N-FET stack circuits described above. Forexample, Vdrive 818 may be negative with respect to reference 802 inFIG. 8 if all FETs are P-channel, and the zener DZ 822 connection isreversed (anode and cathode exchanged).

FIG. 9 is an exemplary circuit that employs both an N-channel FET stackcomprising N-channel FETs M_(N1) 902, M_(N2) 904 and M_(N3) 906, plus aP-channel FET stack comprising P-channel FETs M_(P1) 908, M_(P2) 910 andM_(P3) 912. For Class D operation, an input square wave may be providedwith respect to common 914 at the input 916 to the N-FET stack, andcoupled to an input for the P-FET stack on the gate of M_(P3) 912 via acapacitor C_(GP1) 918. A bias voltage, set for example to one half ofV_(GS)(on) below the P-FET stack reference V_(DD) 930, may be providedfor M_(P1) 908 via a bias resistor R_(BP1) 920.

Alternatively, the capacitor C_(GP1) 918 and the bias resistor R_(BP1)920 may be deleted, and the input 916 and the gate of M_(P1) 908 mayeach be driven, instead, by means of an appropriate non-overlap clockgenerator (not shown).

Control of the N-FETs M_(N2) 904 and M_(N3) 906 is substantially asdescribed with respect to FIG. 2 (for N=3). The gate of M_(N2) 904 iscoupled to common (i.e., decoupled) via a capacitor C_(GN2) 922 having arelatively large value, and may be biased to about (V_(DD)/3) volts viaa bias resistor R_(BN2) 924. The gate of M_(N3) 906 is decoupled tocommon via a capacitor C_(GN3) 926 having a value calculated asdescribed with respect to FIG. 2, and may be biased to (V_(DD)/2) voltsvia a bias resistor R_(BN3) 928.

The P-FET stack is controlled analogously as the N-FET stack. Thepolarities of the bias voltages are inverted, and referenced to the“reference voltage” of the P-FET stack, which in this case is V_(DD)930. For purposes of capacitively decoupling the P-FET gates, the factthat the P-FET reference voltage is V_(DD) 930 is likely to make littledifference, because V_(DD) is typically closely coupled to the circuitcommon 914 that is the reference for the N-FETs. Therefore, decouplingcapacitors 932 and 936 may alternatively be connected to circuit common914. As shown, however, the gate of M_(P2) 910 is decoupled to V_(DD)via a relatively large capacitor C_(GP2) 932, and biased to about ⅔V_(DD) via a bias resistor R_(BP2) 934. The gate of M_(P3) 912 isdecoupled to V_(DD) via a capacitor C_(GP3) 936. The value of C_(GP3)936 may be calculated as described with respect to FIG. 2 for X=3 andN=3. The gate of M_(P3) 912 is biased to about V_(DD)/2 via a biasresistor R_(BP3) 938.

An output voltage Vdrive 940 will be driven between common and V_(DD),according to whether the N-FET stack is conducting or the P-FET stack isconducting. The output Vdrive 940 may be shunt filtered by a shuntfilter 950, and may be processed by a matching and coupling circuit 960,as described below in more detail with respect to FIG. 10. From thematching and coupling circuit 960 the signal may proceed to an antenna942, typically via a transmission line, one or more further filtersections, and an RF switch (not shown).

The shunt filter 950 of FIG. 9 may be similar to that shown in FIG. 4,or that shown in FIG. 6. The matching and coupling circuit 960 of FIG. 9may, for example, be similar to that shown in FIG. 3. However, FIG. 10illustrates filtering that may be employed for both blocks 950 and 960in the circuit of FIG. 9. The capacitor C_(S) 952 may serve as the shuntfilter 950. The remainder of FIG. 10 may function as the matching andcoupling circuit 960 of FIG. 9. An inductor L_(C) 954 may comprise aphysical coupling connection. A coupling capacitor C_(C) 962 serves toblock DC. L_(A) 964, C_(A) 966, L_(B) 968 and C_(B) 970 may beconfigured for matching to the output impedance Z_(OUT), which istypically 50 ohms

VII. Monolithically Integrated, Medium Power Dual-Band RF Transceiver

An RF transceiver, such as the dual-band RF transceiver represented inFIG. 12, typically includes a received-signal amplifier such as items1226 or 1256 of FIG. 12. Such a received-signal amplifier is typically alow noise amplifier (LNA), and is employed to condition signals receivedfrom the antenna. An RF front end may be considered to be an RFtransceiver circuit that does not necessarily include an LNA.

In most RF transceivers, discrete integrated circuits must be combinedin a module to fabricate a complete RF front-end section. Typically, atleast the antenna switch will be fabricated on a different, separateintegrated circuit (IC) from the PA, and often many more discreteintegrated circuits must be connected via off-chip wiring to fabricatean RF front end module. Each such discrete integrated circuit must bedefined by particular performance requirements which ensure that themodule functions properly even when the discrete integrated circuitswhich it comprises are from different lots, or have been designed andmanufactured differently from other integrated circuits that perform thesame tasks. Such performance requirements, which are thus developed toachieve mix-and-match flexibility and reliability, may well exact a costfor the discrete ICs that are combined in these devices.

PAs in multiple-IC transceiver modules typically produce a signal ofsubstantial power on demand. An antenna switch unit couples an antenna(more precisely, an antenna connection) to either a transmit signalmatched to the expected antenna impedance (e.g., 50 ohms), or to areceive signal input. However, damage to the antenna connection or theantenna may cause the impedance reflected to the antenna connectionpoint from the antenna connecting line to vary drastically from itsexpected value. In such event, a large voltage standing wave (VSW) maybe caused by the resulting mismatch between that reflected impedance,and the expected value to which the transmit signal has been matched.Voltage excursions much larger than those expected during normaloperation may be generated as a consequence of such mismatch-inducedVSWs. Voltage withstand requirements for antenna switches are typicallyset much higher than normal peak operating voltages to avoid damageunder such mismatch conditions.

The IC area occupied by switching devices (such as FETs) in apower-switching circuit, such as an antenna switch, may increase as thesquare of the voltage they are capable of withstanding. Thus, halvingthe required withstand voltage may reduce the switch device area to onefourth. Moreover, because these devices dominate the IC area used by anantenna switching circuit, a very substantial saving in IC area (andthus in manufacturing cost) may be realized by reducing their requiredwithstand voltage. Such reduction may not be practical when discrete ICsmust be coupled to fabricate an entire transceiver. However, a single ICthat includes all devices from a PA, through an antenna switch, and toan antenna connection, may take advantage of reliable internal couplingand close device matching to protect against high mismatch-induced VSWs.Due to these advantages of integration, substantial savings in devicearea can be realized as compared to combining discrete ICs to fabricatea comparably-performing transceiver.

FIG. 12 is a simplified block schematic diagram of the primary RFsections of a dual-band transceiver that is configured to benefit fromsuch internal protection. A low-level signal at a first operatingfrequency f_(O1) is coupled to an input node 1202 from a source whichmay (but need not) be on the same IC chip. The signal is amplifiedthrough any suitable amplifier, as indicated by amplifier 1204. Thesignal produced by the amplifier 1204 may deviate considerably from apreferred rectangular shape if a pulse adjustment circuit 1500 isprovided to improve rectangularity, and preferably to also adjust theduty cycle, of the waveform that is ultimately coupled into a poweramplifier (PA) 1206.

The output of the pulse adjustment circuit 1500 is the input to the PA1206, which draws power from a supply V_(DD) 114 via supply conditioningelements, including a series regulator 1400 and an RF choke (RFC) L_(S)108, to generate a PA output signal. The PA output signal has acharacteristic impedance resulting from the input signal, the PA circuitelements, and the supply conditioning elements, and generally differsfrom the impedance expected at the antenna node 1214. A coupling,matching and filtering network may be needed, for example as representedby a block 1210. Such a network may couple the PA output signal to theantenna switch while blocking DC current, and may transform the PAoutput impedance to the desired antenna node impedance (e.g., 50 ohms).It may also filter undesirable signal components, such as harmonics off_(O1), from the PA output signal before coupling it to an “A” input ofan antenna switch 1700. If separate grounds are maintained as a matterof design preference, then the output of the coupling, matching andfiltering block 1210 may be referenced to a ground reference 112, whichmay be distinguishable from a circuit common reference 104 usedelsewhere in the circuit. The antenna switch 1700 selectably couples thesignal to the antenna node 1214, from whence it may be coupled, forexample by transmission line, to an antenna that may be separated fromthe IC chip.

Availability of the antenna connection of the antenna switch on the sameIC chip as the PA (and all intervening circuitry) provides anopportunity to reliably limit the maximum electrical stress that must beendured by the antenna switch circuitry, by the PA, or by coupling,matching or filtering elements. An output sensor 1600 may be coupled tothe antenna node 1214, sensing the electrical stress and providing asignal that will cause the PA to reduce its output if the electricalstresses are excessive. To this end, the output 1220 of output sensor1600 is coupled to an input “B” of a PA control block 1300. An “A” input1224 to the PA control block 1300 may receive an amplitude controlsignal to adjust the envelope amplitude of the PA output signal. Thisinput may also be used to restrict, or even to terminate, output fromthe PA. Both the “A” and “B” inputs may affect an output “D” that iscoupled from the PA control block 1300 to the series regulator block1400. A “C” input 1222 to the PA control block 1300 may be provided withinformation, or a signal, that controls an “E” output from the block1300. The “E” output may be coupled to the pulse adjustment circuit 1500to control the duty cycle of the rectangular wave that is input to thePA 1206. Duty cycle control may, for example, provide another means toreduce the power level of the PA output signal. The signal path from1202 may be tuned for a first band of operating frequencies, whichinclude f_(O1).

The antenna switch 1700 may selectably decouple the antenna node 1214from the first-band transmit signal on input “A,” and couple the antennanode instead to output “B” so as to deliver a signal received from theantenna to a receive preamplifier 1226 for a first receive band. Thereceive preamplifier 1226 (as well as 1256) is preferably a low noiseamplifier (LNA). LNAs are not necessarily included in integrated frontends as described herein, though they typically are included in completetransceiver circuits. The output from receive preamplifier 1226, ifpresent, may be delivered to further first receive band circuitry eitheron or off the same IC chip. The antenna switch 1700 may similarlyselectably couple the antenna node 1214 to a second receive bandpreamplifier 1256 to amplify a signal from the antenna to a secondreceive band output node 1258. That output may be delivered to furthersecond receive band circuitry either on or off the IC chip.

Similarly as described above with respect to the first transmit bandcircuitry, a transmit signal at a second operating frequency f_(O2) in asecond operating frequency band may be provided to an input 1232, andamplified by an amplifier 1234. The duty cycle and waveform of thesignal output from the amplifier 1234 may be conditioned by a pulseadjustment 1501 under control of a PA control block 1301, and thendelivered as an input to a second band PA 1207. The second band PA 1207will generate a second-band PA output signal using power that isprovided from V_(DD) 114, as limited by a series regulator 1401 undercontrol of the PA control block 1301, via an RF choke 109. Thesecond-band PA output will have a characteristic impedance, and will becoupled to the “D” input of the antenna switch 1700 via a block 1211that couples the signal, matches the PA output and antenna nodeimpedances, and filters the output signal. The antenna switch 1700 maybe controlled to couple the “D” input to the antenna node 1214, fromwhence the signal will be delivered to the antenna 1216. The output 1220of the output sensor 1600 may be coupled also to a “B” input to thesecond-band PA control block 1301, whereby excess output voltage willcause the second-band PA output signal to be reduced to safe levels. Thesecond-band PA control block 1301 may also accept an envelope-controlsignal at an “A” input 1254, as well as a duty-cycle control signal at a“C” input 1252.

Though not shown, control circuitry is preferably enabled only when theassociated PA is active. Exemplary circuitry for a PA control block,such as 1300 or 1301, is shown in FIG. 13. Exemplary circuitry for aseries regulator, such as 1400 or 1401, is shown in FIG. 14. Exemplarycircuitry for a pulse adjustment circuit, such as 1500 or 1501, is shownin FIG. 15. The PAs 1206 or 1207 may be fabricated as described forDriver Elements 200 of FIG. 1, together with a shunt filter, such asblock 400 of FIG. 1 or the appropriate elements of FIG. 11. Coupling,matching and filtering circuits 1210 and 1211 may be fabricated asdescribed above for FIG. 3, or FIG. 10, or in any other manner to obtainsimilar coupling, matching and filtering effects. Note that if a circuitas shown in FIG. 11 is to be employed, the coupling capacitor 302 willbe disposed before the shunt filter, and the matching and filtering willbe provided subsequently, as in block 990.

FIG. 13 illustrates exemplary circuitry for a PA control block 1300. Anenable input “A” 1302 may be coupled directly to a FET 1304, such thatif input 1302 is approximately ground potential, an output “D” 1306 candraw no current. Conduction into the output “D” 1306 may control the PAoutput power via a series regulator, such as shown in FIG. 14, such thatwhen output 1306 conducts no current, no current will be provided to thePA, reducing output power to zero as discussed below with respect toFIG. 14. FET 1308 is biased by resistors 1310 and 1312, which may haveequal values, a nominal value such as 30-50 kΩ being selected forengineering convenience. This configuration protects low-voltage FETs,ensuring that V_(GD), V_(GS), and V_(DS) for all of FETs 1304, 1308 and1314 do not significantly exceed V_(DD)/2.

A power sense input “B” 1316 may be coupled to resistor 1318. Resistor1318 may be about 30-50 kΩ, and reasonably equal to a resistor 1320 toestablish unity gain for op amp 1322. A power set input “C” 1324 may beset, in one embodiment, from 0V to 2*Vth (FET threshold voltage), whereVth may be 0.4 to 0.7V, nominally about 0.5V, and is consistent withinthe circuit. The noninverting input of op amp 1322 is prevented fromexceeding this voltage range by means of a resistor 1326 (e.g., 30-50kΩ) together with diode-connected FETs 1328 and 1330, thus limiting themaximum power that may be selected. The skilled person may adjustcircuit values, and circuit design, so as to achieve a selectable outputpower up to a fixed circuit design maximum. In particular, one or bothdiode-connected FETs 1328 and 1330 may be replaced by a network thatincludes a bandgap reference, for example to increase accuracy of powersettings and output voltage limits Many other techniques may be employedto achieve similar effects. When power sense input “B” 1316 exceeds avalue established by the power set input voltage, FET 1314 will ceaseconducting, precluding conduction into output “D” 1306.

The PA control block 1300 also provides an output “E” 1512 to controlthe duty cycle adjustment effected by the pulse adjustment circuit 1500of FIG. 12. A reference voltage, which may be adjustable according tofactors such as fabrication process parameters, is provided at an input1332. This voltage is doubled by an op amp 1334 under control ofequal-valued resistors 1316 and 1318. Of course, in other embodimentsthe gain of circuitry such as shown in FIG. 13 will likely be different,and resistors setting such gains, for example 1316 and 1318, willaccordingly differ in value. By reducing the duty cycle somewhat, the PAoutput power may be correspondingly reduced, and by reducing it to zerothe PA output may be suppressed entirely. A reference voltage providedto an input 1322 of an amplifier 1320, the gain of which may becontrolled in the usual manner by resistors 1324 and 1326, may serve toestablish the voltage at output “E” 1512 to control duty cycle of theoutput of block 1500.

FIG. 14 illustrates an exemplary series regulator circuit 1400 forlimiting the effective voltage provided to the PA, and thus limiting thePA output amplitude. The voltage provided to input 1306, as compared toV_(DD), is divided via resistors 1406 and 1404 to control a P-channelFET 1402 and also protect FET 1402 from excessive voltage between anytwo nodes. A P-channel FET 1408 is biased by resistors 1410 and 1412 soas to divide maximum voltages that are generated between V_(DD) and anoutput 1414, somewhat equally between FETs 1402 and 1408. The output1414 provides power to the PA via an RF choke. The FETs 1402 and 1408may, for example, have V_(GS) threshold voltages of between −0.4V and−0.7V. Resistors 1404, 1406, 1410 and 1412 may all be substantiallyequal, with a magnitude selected for engineering convenience to be, forexample, 30-50 kΩ. These exemplary values and relative values may bevaried for engineering convenience.

FIG. 15 is a schematic representation of an exemplary signalconditioning circuit 1500. An input signal may be provided on an inputnode 1502, and coupled via a diode-connected FET 1504 to the input of aninverter 1506. When the input signal voltage, plus the V_(DS) thresholdof FET 1504, is less than the threshold of the inverter 1506, the outputof inverter 1508 will be low. However, when the voltage of the inputsignal rises above this value, the FET 1504 will cease conducting.Thereafter, even if the voltage of input 1502 is quite high (e.g.,V_(DD)), current from the input will be limited by P-channel FET 1510under control of an input voltage 1512. Such current through the FET1510 must charge a capacitor 1514 (which may be a metal-insulator-metalor “MIM” capacitor of about 0.025 to 0.05 pF) until the input toinverter 1506 rises above its switching threshold. At that point theinverters 1506 and 1508 will change state rapidly due to positivefeedback via the capacitor 1514, providing square edges at an outputnode 1516, rise and fall times being limited primarily by delays throughthe inverters 1506 and 1508. If the signal at the input 1502 is aroughly rectangular wave of about 50% duty cycle, as in a preferredembodiment, the voltage at the control input 1512 may be adjusted suchthat the output duty cycle is reduced from 50% to an arbitrarily lowervalue that may reach zero. The input signal may be configured to have aduty cycle exceeding 50% if a wider range of output duty cycle isdesired.

FIG. 16 is a schematic of exemplary circuitry 1600 for sensing peakvoltage at a sense node 1602, which may for example be connecteddirectly to antenna node 1214 of FIG. 12. An input divider may be usedas shown to sense relatively high voltages. Four roughly equal resistors1604, 1606, 1608 and 1610 having a relatively low resistance, such as 1kΩ, may be used. Diode-connected FET 1612 conducts when this voltage ishigh, providing current through a resistor 1614 of about 24 kΩ to acapacitor 1616 of about 1 pF. Many other values may be used, so long asthe time constant established by resistor 1614 and capacitor 1616 ismuch smaller than the duration of any event that could cause the outputvoltage to rise (for a given level of PA output signal). For example, anantenna mechanical event that caused the antenna impedance to varydrastically from the design value, thereby causing a high voltagestanding wave to appear, will take at least milliseconds to occur. Thetime constant of approximately 24 nS of the exemplary circuit 1600 iswell below such an event duration. However, the corner frequency due tothese components should generally be well below both the first andsecond band operating frequencies f_(O1) and f_(O2), in order to avoidcircuit oscillations. If event durations may approach 1/f_(O), thenother common circuit design considerations may require a morecomplicated circuit to avoid oscillation while ensuring that theresponse is sufficiently fast to prevent excessive voltages.

FIG. 17 is a simplified schematic of exemplary circuitry 1700 for anantenna switch. Further details regarding design and fabrication of suchan RF switch may be found in U.S. Pat. No. 6,804,502, issued Oct. 12,2004 and entitled “Switch Circuit and Method of Switching RadioFrequency Signals.” Circuitry to provide control signals is not shown.Moreover, the control voltages should preferably be either “high,”nearly V_(DD), or “low” at nearly −V_(DD). To generate −V_(DD) controlvoltages, a negative voltage generator will be helpful, preferably alow-noise circuit such as described in copending, published U.S. patentapplication Ser. No. 10/658,154, filed Sep. 8, 2003 and entitled “LowNoise Charge Pump Method and Apparatus.” Such a low-noise charge pump isimportant for avoiding unintended emissions from the antenna.

A port node 1780 is the common connection of the switch 1700. In FIG.12, the common connection of the switch 1700 is coupled to the antennanode 1214. The common connection is generally coupled to only one RFport (port A 1710, port B 1730, port C 1750, or port D 1770) at a time.Each RF port has a corresponding “+” control node and a corresponding“−” control node. For ports A, B, C and D, the “+” control nodes arenodes 1708, 1728, 1748 and 1768, respectively, while the “−” controlnodes are nodes 1718, 1738, 1758 and 1778, respectively.

To couple an RF port to the common connection, a “high” voltage(˜V_(DD)) is applied to the port's corresponding “+” control node, whilea “low” voltage (˜−V_(DD)) is applied to the port's corresponding “−”control node. Meanwhile, a “low” voltage is applied to each “+” controlnode corresponding to another RF port, and a “high” voltage is appliedto each “−” control node corresponding to another RF port. Thereby, aselected RF port will be coupled to the common connection, while everyother RF port will be coupled to ground. Thus, to couple RF port A 1710to common connection 1780, a “high” voltage is applied to control nodes1708, 1738, 1758, and 1778, while a “low” voltage is applied to allother control nodes (1718, 1728, 1748 and 1768).

Every resistor will typically have the same value. In some embodiments,the value will be roughly 30-50 kΩ. The resistor is selected such thatthe time constant of the parasitic gate capacitance of a FET (e.g. M1_(A) 1701), in conjunction with the value of its corresponding gateresistor (e.g. 1704) is much greater than 1/f_(O), where f_(O) is thelowest significant frequency of the RF signal being controlled. Theillustrated configuration serves to divide the voltage appearing acrossFET stacks (such as the stack consisting of FETs M1 _(A), 1701, M1 _(B)1702 and M1 _(C) 1703, the stack consisting of FETs M2 _(A), 1704, M2_(B) 1705 and M2 _(C) 1706, and so on) uniformly, reducing compressioneffects. The FET stacks (such as FETs 1701, 1702 and 1703) that providethe switching functions may include more or less than the three devicesthat are shown for illustration; stacks of at least nine devices havebeen successfully fabricated. Due to the voltage stress distributionuniformity, a wide range of signal voltages and fabrication processparameters may be accommodated.

Integrated Circuit Fabrication and Design

Integrated circuit fabrication details are not provided in the abovedescription. In some preferred embodiments, including some which haveoutput powers in excess of 1 W at around 2.4 GHz, the integratedcircuits may be fabricated in accordance with ultrathin silicon onsapphire processing as described in U.S. Pat. No. 5,663,570, issued Sep.2, 1997 and entitled “High-Frequency Wireless Communication System on aSingle Ultrathin Silicon On Sapphire Chip.” Othersemiconductor-on-insulator (SOI) techniques may be used to fabricate adual-band transceiver integrated circuit as described above, for atleast some frequency bands and power levels.

The preferred integrated circuit fabrication techniques described abovereadily produce FETs having a rather low maximum V_(DS). Accordingly,various techniques are described for stacking FETs to achieve control ofhigher voltages while maintaining consistent processing. Using othermanufacturing techniques, or lower voltages and impedances, a need forcascode or multiply-stacked FETs may be avoidable.

Conclusion

The foregoing description illustrates exemplary implementations, andnovel features, of a method and apparatus that employs stackedtransistors to control conduction between a pair of nodes in anintegrated circuit. The skilled person will understand that variousomissions, substitutions, and changes in the form and details of themethods and apparatus illustrated may be made without departing from thescope of the invention. Numerous alternative implementations have beendescribed, but it is impractical to list all embodiments explicitly. Assuch, each practical combination of the apparatus or method alternativesthat are set forth above, and/or are shown in the attached figures,constitutes a distinct alternative embodiment of the subject apparatusor methods. Each practical combination of equivalents of such apparatusor method alternatives also constitutes a distinct alternativeembodiment of the subject apparatus or methods. Therefore, the scope ofthe presented invention should be determined only by reference to theappended claims, and is not to be limited by features illustrated in theforegoing description except insofar as such limitation is recited, orintentionally implicated, in an appended claim.

It will be understood that similar advantages of integration will accrueto circuits having other functional blocks. For example, mixers may beincorporated on such a device, enabling integration of more portions oftransmission signal processing. Phase locked loops may further enhancethe ability to generate the transmission signal on the same monolithicIC as the RF front end or transceiver. Additional types of filters maybe useful, for either or both of receive and transmission processing.

All variations coming within the meaning and range of equivalency of thevarious claim elements are embraced within the scope of thecorresponding claim. Each claim set forth below is intended to encompassany system or method that differs only insubstantially from the literallanguage of such claim, if such system or method is not an embodiment ofthe prior art. To this end, each described element in each claim shouldbe construed as broadly as possible, and moreover should be understoodto encompass any equivalent to such element insofar as possible withoutalso encompassing the prior art.

What is claimed is:
 1. An integrated RF Power Amplifier (PA) circuit,comprising: a) an input node to accept an input signal with respect to areference voltage Vref, coupled to a gate G1 of a first insulated-gateFET M1; b) a plurality of additional insulated-gate FETs M2 to Mn havinga same polarity as M1 and coupled in series with M1 to form a controlcircuit configured to control conduction between the reference voltageand an output drive node, wherein FETs M2 to Mn are each enslaved to M1;c) an output coupling capacitor coupling the output drive node to anoutput load node; and d) an output power control input node coupled to apower controlling insulated-gate FET that is coupled in seriesconnection with M1.
 2. The integrated circuit of claim 1, wherein thepower controlling FET is disposed between the reference voltage and apower supply voltage external to the PA.
 3. The integrated circuit ofclaim 1, wherein the power controlling FET is one of the FETs M2 to Mn.4. The integrated circuit of claim 1, further comprising a low-passfilter disposed in series with the output coupling capacitor between theoutput drive node and the output load node.
 5. An integrated RF PowerAmplifier (PA) circuit, comprising: a) an input node to accept an inputsignal with respect to a reference voltage Vref, coupled to a gate G1 ofa first insulated-gate FET M1; b) a plurality of additionalinsulated-gate FETs M2 to Mn having corresponding gates G2 to Gn and asame polarity as M1 and coupled in series with M1 to form a controlcircuit configured to control conduction between the reference voltageand an output drive node, wherein FETs M2 to Mn are each enslaved to M1;c) an output coupling capacitor coupling the output drive node to anoutput load node; and d) a corresponding predominantly capacitiveelement connected directly between each gate, G2 to Gn, and Vref.
 6. Theintegrated circuit of claim 5, further comprising (e) an output powercontrol input node coupled to a power controlling insulated-gate FETthat is coupled in series connection with M1.
 7. The integrated circuitof claim 6, wherein the power controlling FET is disposed between thereference voltage and a power supply voltage external to the PA.
 8. Theintegrated circuit of claim 6, wherein the power controlling FET is oneof the FETs M2 to Mn.
 9. The integrated circuit of claim 5, furthercomprising a low-pass filter disposed in series with the output couplingcapacitor between the output drive node and the output load node.